Testing ERA-5 MMIC for use as a multiplier

An easy way to generate harmonics is to drive an amplifier into compression. MMIC amps are easy to use and the 1 DB compression point is listed in the datasheet for just about all MMIC amplifiers available today. Normally this would be an undesired operating point. If you need a multiplier this a simple way to do it. Output are typically higher using an MMIC as a multiplier than using a simple diode multiplier. The output circuit of the MMIC can be designed to enhance the desired harmonic. That was not done in this test.

ERA-5 gain below 1GHZ = 20 DB
1 DB compression = +17.5 DBm

Input 738 MHz into MMIC at +6 DBm and +2 DBm drives the amp well into compression. No attempt was made to optimize the output for a particular harmonic or to suppress the fundamental. The output capacitor was 12 pf to favor the upper frequencies. Results varied depending on the input frequency. The data below is for 738 MHz. At 571 MHz the 5th harmonic with +2 DBm input was -20 but had a lot of “grass” around the carrier. Lowering the input to +1 DBm dropped the harmonic to -30 DBm and cleaned up the noise (or it dropped below the noise floor of the spectrum analyzer).

It appears the input should not exceed 5 DB above the 1 DB compression level or excessive noise will result.

+6 DBm input is 8.5 DB above the 1DB compression point
+2 DBm input is 4.5 DB above the 1DB compression point

Output 1 is with a +6 DBm input
Output 2 is with a +2 DBm input

Frequency(MHz) or Harmonic/Output1/Output2
738/+20/+18
X2/-2/-6
X3/+3/-12
X4/-4/-30
X5/-20/-30
X6/-42/-34
X7/-38/-46
X8/-30/-42
X9/-40/-50
X10/(below noise)/(below noise)

Oscillators

I assembled the two transponder oscillators yesterday. One is for 82 MHz and the other for 114.2 mHz. They are based on the initial design I did last summer. I thought I had these all figured out. The 82 MHz oscillator just sucked. The harmonic content was very high. Hell, I did not need the multiplier stages I designed. I had a 738 MHz harmonic from the oscillator directly only 30 DB below the fundamental!! As it turned out the little MMIC amp I had at the end of the oscillator was being overdriven. I just removed the MMIC and jumpered the board from the end of the buffer to the SMA connector. I have a good -5 to 0 DBm which was the initial design goal. I also removed the filter components after the buffer. I was getting a spur around 300 MHz. I don't know if it would have gone away by removing the MMIC. I removed the filter parts before removing the MMIC. I suspect circuit board traces in the filter caused the instability. I probably had stray capacitance and/or inductance somewhere in the filter physical layout that screwed it all up.

Other changes I made:
1. Removed the tuned transformer between the oscillator and buffer. Replaced it with a short to the DC line.
2. I moved the input capacitor of the buffer that was on the transformer link to the top of the oscillator emitter resistor.
3. I also reduced the buffer emitter resistor from 220 to 100 ohms. This helped reduce the harmonics a little. I think the buffer was also being overdriven.

I experimented with the tapped capacitors in the oscillator tank circuit. Increased them to 82 and 33 from 68 and 22 pf. No real difference. The oscillator may be slightly more stable, from a start up perspective, with the original caps so I put them back to the way they were.

In the end the 2nd harmonic is 10 db down and the others are 30 db down or more. The 5th harmonic is the last one I see on the spectrum analyzer. The input tuning on the multiplier may make these spurs a moot point. If not I will have to add an extra low pass filter after the oscillator. The final oscillator is much simpler design.

I will incorporate the design changes into the 114.2 MHz oscillator and test it out tomorrow night.

I guess I will have to go back and update the design document on my web page. That will have to wait until after Hamvention. I need to get this project done by Hamvention.

Interdigital Filters - remade

I etched the new boards over the weekend. The frequency of the 738 and 571 filters came out spot on. The pesky "step" is still there especially on the 571 filter. The loss is less on the 571 but that was expected per the simulation using FR4 board stock. The return loss just sucked on both. I ended up adding a surface mount trimmer cap (3 to 12 pf I think) in series with the input on the 571 filter. I also moved the input tap up several millimeters. About as far as I could reach with the body of the trimmer. Adjusting the cap I was able to achieve a return loss of 18 DB. Slightly above the desired frequency I could get 20. I adjusted for best RL on the design frequency. When I get some time I will add the same capacitor to the 738 MHz filter. For now these are done and I will go on to other modules of the project.

Making New Artwork for 738 Interdigital Filter

One final mucking with Ansoft dimensions on the interdigital filter. I started by changing the microstrip impedance to 82 ohms (40 mils on this FR4 board). It was around 40 or 50 before from the initial Ansoft design. That appears to work much better. On the pipe cap filters the ideal impedance of the pipe coax structure was 77 ohms so that must hold in general for the micro strip filters too. The filter response is much better with the best return loss also being the center of the band pass response. The RL looks to be 15 or so. Insertion loss is still high but down under 15 DB now. Out of time for tonight so I will re-do the PC board artwork in Eagle tomorrow night.

Update on Frequency Error in Ansoft Design

I found the source of the 100 MHz frequency error in the earlier post. Ansoft converts the theoretical filter design into a physical layout. This layout is more of a schematic then a circuit board layout. The four microstrips of the filter are modeled in Ansoft as four rectangles. Three sets of these rectangles make up the entire filter. Each strip is broken up into three pieces because of the input and output taps. The taps are represented by a “T” symbol that has no length, just width. To position the tap the total length of the line must be broken up in the model representation. There is some confusion as to what length “p” is in relation to the total length of the line. Also the conversion process did not use the variable “p” but instead inserted the total length of the line. When I made the board I misinterpreted the meaning of each length “p”, i.e. p1, p2 and p3. It was not clear how these added up to make the total length. I ended up with each filter element being about 250 mils too long. This made the filter resonate 100 MHz low. Now that I have figured this out the simulated frequency and measured frequency is within reason give just an uncertain dielectric constant of FR4 at these frequencies.

Measurements on the N4 Interdigital Filter

After messing around with this interdigital filter I never could get a nice smooth response. At best there is “step” in the response as the frequency goes above the center frequency of the pass band. For this application that would not be a problem as I am just removing other harmonics of the main oscillator. Using FR4 board is not the best for UHF and above filters. The loss is just too high. Also the dielectric constant is not very constant. Now that I have this filter built I measured the final microstrip elements.

P (total length of each microstrip, there are four) = 2062.5 mils
P2 (tap for the input and output from ground end) = 203 mils
W (width of each microstrip, they are all the same) = 164 mils
S1 (spacing between element 1 and 2) = 391 mils
S2 (spacing between element 2 and 3) = 172 mils
S3 (spacing between element 3 and 4) = 500 mils

Measuring the actual center of the band pass on the tracking generator indicates 737.68 MHz. Now we can modify the simulation with our actual measured values and keep changing the dielectric constant until the frequency matches the measured. Nominally FR4 is around 4.5. The simulation shows:

For 4.7 Fo = 743.53 MHz
For 4.8 Fo = 736.39 MHz
For 4.9 Fo = 728.57 MHz
For 4.79 Fo = 737.68 MHz

Setting the simulation to 4.79 yields our measured frequency of 737.68. We will use this value for future filters around this frequency. I want to try a hairpin filter next. I have the initial design. Other than the narrow spacing and I assume more critical layout tolerances the pass band does not contain the “step”. Hairpins do respond to the even harmonics of the first pass band. I don’t think that will be a problem the higher harmonics of the multiplier will be reduced by the design of the multiplier.

Documenting the shortening of the MicroStrips

Start length = 61mm

 

Pass band frequency per length of each microstrip:

61mm = 630 MHz

60mm = 642.5 MHz

59mm = 657 MHz

57mm = 678 MHz

54mm = 715 MHz

53mm = 728 MHz

52mm = 736.7 MHz

 

 

After Microstrip Adjustments

I have made the modifications to the 738 filter PC board layout.  I shortened the strips to bring the center frequency to 738 MHz. The original length from the Ansoft calculations was 61mm.  I had to shorten these to 52mm.  That is a reduction of 14.75%.  My guess the reduction is required because of stray capacitance on the board.   The final 3DB bandwidth comes out to 15.1 MHz.  The final gain with the +20 DB MMIC is 0DB.  My insertion loss is 20DB behind that amp.  This is in line with Ansoft’s calculations using FR4 board.  Rogers PC board would be a better choice because of the dielectric loss tangent (TAND).  It is 0.02 on FR4 but only 0.0009 for Duroid 5880.

 

The input return loss is interesting.  Using FR4’s TAND value Ansoft shows the best return loss as only 10DB but above Fo at 756 MHz.  This is less than the original 18DB I had before mucking with the strips.  I see this on my tracking generator.  I was able to improve this to 15DB by adjusting the tap point on the first microstrip element.  I moved it a millimeter or two toward ground.  I don’t know that it improved return loss at 738 as that still looks to be 10DB. 

 

 

 

738 MHz Bandpass Microstrip tuning

I was playing around with Ansoft this morning and the 738 filter. First I adjusted the dielectric constant to see if I could account for the lower filter frequency response.  I found the dielectric constant would have to be something like 7+ to drop the frequency down 100 MHz.  Design spec for FR4 at lower frequencies is 4.5.  I don’t think 7 is a reasonable value even at 700 MHz.   I then checked my actual dimensions of the micro strip.  They are within 0.5 millimeter of the design values.  Again, not enough to account for 100 MHz.  I am beginning to wonder if stray capacitance may be lowering the frequency. 

 

Another finding was on the insertion loss.  When I modeled this board I did not set the dielectric loss tangent value.  I see from documentation that this should be 0.02 for FR4 board.  Doing that I can now account for the loss.  The insertion loss is 18 dog biscuits (Jake is pawing me as he wants some of those).  That agrees with my board.  My MMIC amp on the output has a +20 gain at 700 MHz.  That gives me the +2 DB output with a 0-DBm input signal I see on the analyzer.   Using Rogers 5880 the TAND values is 0.0009 which would produce a more reasonable loss of 4 dog biscuits and improve the return loss.   

 

Next I will shorten the strips to see if I can bring the frequency up to 738.  I can try to use the same technique I used for the pipe filters to tune each stage by lightly coupling a spectrum analyzer to the first stage then short out the other stages as I tune down the line.

 

 

First Microstrip Bandpass Filter Results

I etched the board for the microstrip band pass filter.  The desired center frequency is 738 MHz.  The problem with designing such a filter on FR4 PC board stock is the variation of the dielectric constant at the desired frequency.  The second source of errors in the layout of the board itself.  At UHF and above it is difficult to get the necessary precision where a quarter of a millimeter could make a big difference.  My results of the first filter look promising.  The filter response was about what I wanted except the frequency was off by 100 MHz.  The center of the bandpass was 630 MHz and not 738.  I think the insertion loss may be high but it is masked by the output MMIC amp I have on the board.  The overall gain is +2 DB because of the amplifier.  While this does not sound bad remember that the MSA-0886 amp has about 20 DB of gain at 600 MHz. 

 

The plot of the filter response shows a “step” on the upper frequency side of the curve.  This indicates one of the stages is not aligned on the same frequency as the others. 

 

Anyway, the initial measurements:

 

Input return loss at center of bandpass:  18 DB

Overall gain with MMIC amp: +2 DB

Center frequency of band pass:  630.68 MHz

3db bandwidth:  638.33 – 623.64 MHz = 14.7 MHz

 

The filter was designed using Ansoft Designer SV (student version). I will now go back to the design and modify the FR4 dielectric constant to see if I can account for the 100 MHz frequency error.  The idea is to use the modified constant to rework the board dimensions to see if I can get closer to 738 MHz and improve the insertion loss. 

 

As for this board it has served it purpose.  I can try to shorten the microstrips and use capacitors to tune the sections.  I can not adjust the coupling other than to solder metal tabs onto the microstrips and use them to increase coupling to the next strip. 

 

 

 

 

Tinkering With That Darn Oscillator

It appears the high harmonic output is generated by the last MMIC amp.  Not sure why unless it is sensitive to the impedance seen by the input looking back into the filter.  There is a simple low pass filter after the oscillator buffer.  That filter should have been after the MMIC amp but the amp was an after thought when I designed the PC board.  The harmonics go away when I remove the last output capacitor from the filter.  I then have the second harmonic down 35 DB.  I attempted to replace the first and last capacitor with trimmers to see if I could improve the situation.  Not really, so I left it as it was minus the last capacitor. 

 

I adjusted the oscillator transistor bias resistors to improve the output while maintaining the same 35 db harmonic attenuation.  The oscillator output level is now zero DBm.

 

That still puzzles me as to why I am getting the harmonics.   The MMIC is not being overdriven.  The MMIC is an Agilent ABA-53563 I got from Mouser Electronics.  The 1-DB compression point is +14 DBm output.  I appear to have around -15 DBm going into the MMIC so with a 21.5 DB gain that is below the 1-DB compression point.  I checked my spectrum analyzer calibration against my signal generator.  The spectrum analyzer reading agree with the settings of the generator. 

 

I will leave the oscillator as it is and get back to testing the frequency calibration and oscillator heater.  Something tells me I will be redesigning the PC board moving the filter after the MMIC. 

Oscillator Oven

I have been working on a few oscillators for a radio project.  The test oscillator is crystal controlled on a carrier frequency of 104.727270 MHz.  This will be multiplied up to a higher frequency.  Unfortunately I forgot what oven temperature I ordered the crystal for.  I think it was around 50C.  I did not want it to be too high as I am using Styrofoam 3/4” thick for the insulating material. The entire oscillator is heated.  The oscillator is a Butler design which uses a series resonant crystal.  The only tuning is from the resonant circuit of the Butler design.  I set the trimmer capacitor for the best oscillation then adjusted the oven heater until I moved it to the marked crystal frequency.   The oscillator has a temperature sensor on it so I can monitor the temperature.  This makes adjusting the heater easier.  The sensor outputs 0.01 volts/degree K. 

 

I now have a stable operation at a sensor reading of 3.292 V which equates to around 56C.  The frequency counter reads 104.727284 MHz.  The counter is locked to 10-Mhz GPS disciplined oscillator to ensure a correct readout.  I am 14-hz off frequency at this point.  The crystal changes about 57-Hz for a 2 degree C temperature change.  This is around 0.54 PPM.  The oven heater resting current is 0.06 Amps.  The heater is nothing more than a power FET attached to the side of the aluminum oscillator housing.  A thermistor on the oscillator PC board in the gate circuit controls the FET.  An external pot in series with the thermistor is used to adjust the temperature.

 

Now I will remember to write down the temperature specification when I order any new crystals.  55 degrees C. works out well for the type of heater and insulating material I am using. 

 

The next test will be to see how well the heater can hold this temperature when I subject the oscillator package to a temperature change.  I need measure the external and internal temperature while the environment changes.  I can do this by using the GPIB interface on my DVM for the on-board sensor and the RS-232 interface on my Radio Shack hand held DVM.  The sensor is a LM324.  I hope to be able to hold +/- 50 hz from room temperature to an ice bath. That would be 2 degrees C temperature variation.